Low-noise flexible frequency clock generation from two fixed-frequency references

ABSTRACT

A number of methods and clock generator units are disclosed to produce low Phase Noise clocks for use in Radio Frequency systems. The methods and clock generator units all use two reference clocks: a frequency-accurate reference that has comparatively high Phase Noise, and a frequency-inaccurate reference such as that from a BAW or MEMS clock source that has comparatively low Phase Noise. By combining multiple Phase-Locked Loops and a mixer, it is possible to produce flexible output frequencies whose frequency accuracy is derived from the first reference clock but whose Phase Noise level is derived from the second reference clock, all in a readily-integrated and relatively low-cost system.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a Continuation of U.S. patent application Ser. No.14/295,742, filed Jun. 4, 2014, which claims the benefit of priority ofU.S. Provisional Patent Application No. 61/831,095 filed Jun. 4, 2013.The contents of U.S. patent application Ser. No. 14/295,742 and of U.S.Provisional Patent Application No. 61/831,095 are hereby incorporated byreference.

FIELD

The present disclosure relates generally to radio frequency (RF)systems. More particularly, the present disclosure relates to systemsand methods of RF systems that require a wide tuning range and low PhaseNoise integrated clock generators.

BACKGROUND

Effects of local oscillator (LO) noise on RF systems:

In many RF systems, the Phase Noise of a LO clock source is a keyconcern in designing the system. An example of quadrature RF receiver isshown in FIG. 1, while an example of a quadrature RF transmitter isshown in FIG. 2.

In the quadrature RF Receiver of FIG. 1, an RF signal is received by anantenna 101, amplified by a Low-Noise Amplifier (LNA) 102, and thendown-converted from RF to an Intermediate Frequency (IF) or Baseband(BB, sometimes also known as “Zero-IF” or ZIF) by mixing an output fromthe LNA 102 with two versions of a LO clock source using mixers 104. Onecopy of the LO clock source is shifted by a quarter phase, illustratedby a phase shifter 105. The outputs from the mixers 104 are thenfiltered by two Low-Pass Filters 106 for further processing. TheQuadrature RF Transmitter of FIG. 2 is qualitatively similar, butoperates in reverse. As shown in FIG. 2, two BB/IF signals areup-converted to RF using two mixers 204, summed together by a summingdevice 203, and finally amplified by a Power Amplifier (PA) 202 fortransmission over an antenna 201. In these Figures, and in allsucceeding figures in this document, an attempt is made to use commonnumbering schemes for common elements for clarity.

There are many alternate forms that RF systems can take beyond what isshown in FIG. 1 and FIG. 2, and the details of their construction arebeyond the scope of this document; however, almost all structuresinvolve mixing signals with LO clock sources to convert BB/IF signals toRF or vice versa. There are many possible mixer structures, and thedetails of their construction are also beyond the scope of thisdocument, however at their core they can all be modeled as analogmultipliers. Applying two sinusoidal inputs to an analog multiplier:i ₁=cos(2·π·f ₁ ·t)i ₂=cos(2·π·f ₂ ·t)  Equation 1

Results in:

$\begin{matrix}{{i_{1} \cdot i_{2}} = {{{\cos( {2 \cdot \pi \cdot f_{1} \cdot t} )} \cdot {\cos( {2 \cdot \pi \cdot f_{2} \cdot t} )}}\mspace{211mu}{Equation}{\mspace{11mu}\;}2}} \\{= {\frac{1}{2} \cdot ( {{\cos( {2 \cdot \pi \cdot ( {f_{1} + f_{2}} ) \cdot t} )} + {\cos( {2 \cdot \pi \cdot ( {f_{1} - f_{2}} ) \cdot t} )}} )}}\end{matrix}$

In other words, the act of multiplying two pure sinusoids results in twoother sinusoids, one at the “sum” frequency (i.e., f₁+f₂), the other atthe “difference” frequency (i.e., f₁−f₂). In many RF applications,typically one of these two frequencies is desired while the other (knownas the “image”) is not and is rejected at the output, using for example,filters or trigonometric identities.

FIGS. 3A and 3B show the Power Spectral Density (PSD) of one possiblefrequency plan for a Single-Frequency Receive RF system and shows theeffect of LO Phase Noise at the output. Before the mixer, at FIG. 3A, anLO clock source 301 and an RF signal 311 are present, while after themixer, FIG. 3B, there are two signals (two signal components), thedesired “difference” signal 313 (frequency difference component) at IFand the undesired “sum” signal 314 (frequency sum component) at a muchhigher frequency, which can be removed with a Low-Pass Filter (e.g., theLow-Pass Filters 106 in FIG. 1). The effects of Phase Noise 302 on theLO clock source 301 (FIG. 3A) are also shown at FIG. 3B, and result inspectral growth of the RF signals 315 and 316 at the mixer output.

The LO Clock Source Phase Noise 302, when looked at on a PSD plot as inFIG. 3A, is usually expressed in the units of decibel per hertz withrespect to carrier power, or dBc/Hz, i.e., the amount of Phase Noisepower present in 1 Hz of bandwidth relative to the power of the LO tone.

In broadband applications such as telecommunication, Phase Noise isoften expressed using terms such as Integrated RMS Jitter or TotalJitter, often measured in femtoseconds (fs) or picoseconds (ps). Howeverin RF applications it is often more appropriate to talk about PhaseNoise in dBc/Hz at certain frequency offsets from the carrier, forexample “−153 dBc/Hz at 800 kHz offset”. To understand why, FIGS. 4A and4B show a frequency plan for a multi-carrier RF Receiver, where thedesired signal 411 is present but at a much lower power than a secondsignal (also known as a “Blocker” signal) at an adjacent carrierfrequency 421. After being subjected to a mixer, the two output signalsfrom the desired RF channel 411 (FIG. 4A) are present as signals 413 and414 (FIG. 4B), as are the two output signals from the Blocker signal 421(FIG. 4A), namely signal 423 and 424 (FIG. 4B).

As before, the Phase Noise 302 (FIG. 4A) on the LO Clock Source resultsin spectral regrowth, however in the multi-carrier scenario the spectralregrowth 425 from the high-power blocker 425 appears in the IF band ofthe down-converted signal 413. Because it is impossible to remove thisnoise from the IF signal, this irreparably harms the Signal-to-NoiseRatio (SNR) of the down-converted signal 413, limiting the availableinformation-carrying bandwidth of the RF Receiver in that channel. Thespectral regrowth 315 (FIG. 3B) of a single-carrier system is typicallymuch less disruptive than in a multi-carrier system because the regrowthpower is proportional to the signal power, whereas in a multi-carriersystem, the regrowth power is proportional to the blocker's signalpower, which, depending upon a number of factors, can be much higher.

Because Multi-Carrier RF Transmit systems are usually dealing withmultiple RF signals of similar power, spectral regrowth concerns due toLO Phase Noise is often less of an issue than in Multi-Carrier RFReceive systems, but should still be considered.

Phase-Locked Loops and Phase Noise:

There is great deal of information known to those skilled in the artdealing with modeling of Phase Noise in general and LO Phase Noise inspecific; however it is beyond the scope of this document to discussthis in great detail. In summary, all electronic components are capableof generating and modifying Phase Noise, with different generation ormodification characteristics depending upon the component. It is,however, appropriate to provide some background on Phase Locked Loopsand Phase Noise.

An elementary Phase Locked Loop (PLL) is shown in FIG. 5, and comprisesa Phase Detector 501, a Loop Filter 502, and a Voltage-ControlledOscillator (VCO) 503. The Phase Detector compares the relative phases ofthe Reference Clock input and the Output Clock from the VCO and, throughnegative feedback, produces a control signal that is in turn filtered bythe Loop Filter to drive the VCO so that the relative phases of the twoclocks are fixed.

The PLL of FIG. 5 has a simple configuration, however it demonstratestwo key features. The first feature is that by locking the relativephases of the two inputs of the Phase Detector 501, the frequency ofthose inputs is also locked, i.e., the Output Clock frequency is thesame as that of the Reference Clock.

The second key feature is that the PLL is a Phase Noise filter: everyPLL has a loop bandwidth set by the characteristics of the loopcomponents and, depending upon where Phase Noise is added in the system,this noise will see either a low-pass or a high-pass filtercharacteristic. For example, Phase Noise at the input to the PhaseDetector 201 (coming either from the Reference Clock or from the OutputClock) will see a low-pass characteristic, whereas noise that arisesfrom the VCO 503 will see a high-pass characteristic. Put another way,the Phase Noise seen at the Output Clock will be divided into two parts:below the loop bandwidth, the Output Clock Phase Noise will track thePhase Noise from the Reference Clock input (the Phase Noise from the PLLVCO component will be attenuated), while above the loop bandwidth, theOutput Clock Phase Noise will track the Phase Noise generated by the VCO503 (Phase Noise from the Reference Clock input will be attenuated).

It is important to note that the Phase Detector 501 is often a hybridblock, known as a Phase/Frequency Detector. During initial acquisitionand while the frequencies at its input are radically different the blockoperates as a Frequency Detector, but when the input frequencies areclose together the block operates as a Phase Detector. This dualoperating mode is used to ensure reliable operation, and whenalternative circuits are used for Phase Detection, provisions must betaken in the design to ensure that the block still locks to the correctfrequency. The details of this are beyond the scope of this document butare well known to those skilled in the art.

Depending upon the required Phase Noise for the application and upon thecomponents available, a system designer will attempt to choose a PLLbandwidth to meet these requirements. The designer is free to set thebandwidth as low as they desire, with practical limitations primarilyarising from component choices for the Loop Filter 502, but the maximumbandwidth is limited by discrete-time effects to some fraction (oftentaken to be 1/10) of the Phase Detector input frequency. Assuming thatthe desire is for a low Phase Noise over a wide frequency range for usein a multi-carrier RF system and that an extremely low Phase NoiseReference Clock were available but the VCO had relatively large PhaseNoise, the system designer would choose the PLL bandwidth to be as highas possible (for example, greater than or equal to the requiredlow-noise frequency range) to suppress as much noise from the VCO aspossible. Alternatively, if the Reference Clock had comparatively highPhase Noise (as is often the case if it is being provided by some sortof network timing system) but an extremely low Phase Noise VCO wereavailable, the designer would choose a low PLL bandwidth (for example,lower (or much lower) than the carrier spacing) in order to suppress asmuch noise from the Reference Clock as possible.

The PLL of FIG. 5 is relatively limited because it passes its inputfrequency to its output with no modification. A more complex PLL, shownin FIG. 6, is capable of transforming one clock frequency to another bythe addition of three dividers: a Feedback Divider 604, a PostscalerDivider 606, and a Prescaler Divider 607.

As before, once the PLL is locked, the inputs of the Phase Detector 501are phase and frequency locked, however now the Output Clock frequencyis determined by Equation 3:

$\begin{matrix}{F_{OUT} = {\frac{F_{REF}}{M} \cdot \frac{N}{P}}} & {{Equation}\mspace{14mu} 3}\end{matrix}$

The Feedback Divider 604 causes the VCO output to be a multipliedversion of the Phase Detector input, while the Postscaler divider 606and the Prescaler divider 607 both act to produce a Final Output Clockwith a frequency that is a rational fraction of the Reference Clockfrequency. Even though the Output Clock frequency is a rational fractionof the Reference Clock frequency, this PLL is commonly referred to as anInteger-N PLL because all dividers are integers. Phase Noise of the PLLof FIG. 6 is similar to that of FIG. 5, in that the Output Phase Noisehas both low-pass and high-pass components, but there are several newconsiderations that must be taken into account. First the addition ofthree dividers has created three new sources of Phase Noise that must beconsidered in the system budget. Second, the Phase Noise due to theReference Clock at the Output is now multiplied by the same factorN/(M·P) as the frequency multiplication of Equation 3, which means thatlow Phase Noise systems should ideally have comparatively lowmultiplication factors. Third, Phase Noise due to the Loop Filter 502and the VCO 503 is attenuated by the Postscaler Divider 606, whichcreates new opportunities for low Phase Noise systems if the desiredfrequencies are comparatively low. Fourth, certain non-idealities in thePhase Detector 501 and Loop filter 502 can create a Phase Noise “spur”at an offset frequency equal to the Phase Detector frequency, whichcreates new constraints in system frequency planning. Finally, theinclusion of the Prescaler 607 divider results in a lower frequencyFref/M at the Phase Detector 501 input, which reduces both the maximumavailable loop bandwidth and brings the Phase Detector spur locationcloser in frequency to the desired output clock.

In flexible RF systems, the frequency spacing at the output from thefrequency-generating PLL is important. For an Integer-N PLL, frequencyspacing at the VCO output is set by the Phase Detector input frequency,F_(REF)/M. Fine frequency spacing implies a comparatively low PhaseDetector input frequency, however that requirement is at odds with whatwould be desirable for low Phase Noise: maximizing F_(REF) whileminimizing the multiplication factor N/(M·P). As a result, most RFsystems use Fractional-N PLL's, instead of Integer-N PLL's.

An example of a Fractional-N PLL is shown at FIG. 7. The Fractional-NPLL replaces the Integer-N PLL fixed Feedback Divider 604 (FIG. 6) witha programmable Feedback Divider 704 and a Fractional-N Modulator 705(FIG. 7). By introducing the programmable Feedback Divider 704 and theFractional-N Modulator 705, the effective feedback divide ratio is givenby Equation 4:

$\begin{matrix}{F_{OUT} = {{\frac{F_{REF}}{M} \cdot \frac{\overset{\_}{X}}{P}} = {\frac{F_{REF}}{M} \cdot \frac{n/d}{P}}}} & {{Equation}\mspace{14mu} 4}\end{matrix}$where X is the average value of X over time and X=n/d, where n and d arethe Numerator and Denominator of a rational fraction and are inputs tothe Fractional-N modulator 705, which is often implemented usingDelta-Sigma (ΔΣ) techniques. The term Fractional-N comes from theinclusion of this rational fraction.

With this modification, the Phase Detector input frequency F_(REF)/M canbe significantly raised to no longer be the required frequency spacing,while at the same time, fine frequency resolution can be achieved byusing large integer values for both n and d. In many practicalFractional-N PLL based Clock Generators, these coefficients are fairlylarge (perhaps 40 bits or more) to give ultra-fine frequency resolution.This increase of the Phase Detector frequency allows for wider PLLbandwidths and allows for the use of higher-frequency references,thereby reducing overall Phase Noise.

The addition of the Fractional-N divider subsystem (704 and 705)increases the Phase Noise seen at the input of the Phase Detector,however this Phase Noise will be attenuated by the PLL's low-pass filtercharacteristic before being seen at the output, and there are severaltechniques well known to those skilled in the art that allow for furtherreduction of low-frequency Phase Noise from this divider, minimizing itsimpact on the output.

Integrated Low Phase Noise LO Clock Generation Challenges:

The challenges of creating low Phase Noise LO clock sources inintegrated forms are several. First and foremost, in many modernmulti-carrier RF systems, low Phase Noise is required over frequencybands that cover extremely wide bandwidths, perhaps as high as 100-200MHz. At the same time, the LO clock generator is required to have veryfine frequency resolution, perhaps as low as 100 kHz. Additionally, thereference clocks available for generating the LO clock source are oftenderived from network timing references, and while they are extremelyaccurate in frequency, have poor Phase Noise characteristics.

Oscillators are fundamentally integrators, integrating frequency overtime to produce phase, and are constructed by coupling an active circuit(an amplifier) to a narrow band-pass filter, often called a “resonator”.The better the Quality factor (also known as “Q” defined as theresonance frequency divided by the resonance width) of the band-passfilter and the lower the noise of the amplifier, the lower theoscillator Phase Noise will be. In an integrated circuit (IC) there aremany oscillator topologies possible, however in general the highest-Qtopologies use an Inductor-Capacitor (LC) tank network as the band-passfilter, and in general most integrated LC oscillators are limited to Qfactors of 20 or so.

In comparison, Quartz Crystal oscillators can have Q factors greaterthan 10000, however they are impossible to fabricate using standard ICtechnologies. Crystal oscillators can be packaged with standard ICs, butthey require specialized packaging techniques including hermeticsealing. In addition, Quartz Crystal oscillator frequencies arecomparatively low (10-100 MHz), and require comparatively largemultiplication factors to use for many modern RF systems. SurfaceAcoustic Wave (SAW) devices (Q factors of 1000 or more) have been usedas resonators in oscillators, however they are physically large and alsorequire specialized packaging and as a result are comparativelyexpensive. Micro Electrical-Mechanical Systems (MEMS) based resonators(Q factors of 1000 or more) are readily fabricated with standard ICprocessing and are relatively inexpensive; however they havecomparatively low oscillation frequencies (10-100 MHz). Bulk AcousticWave (BAW) devices are qualitatively similar to MEMS, have Q factors of500 or more, are relatively easily packaged together with standard ICdevices, are relatively inexpensive, and operate at high frequencies,0.5-3 GHz. Because of their relatively high Q factors and low PhaseNoise, relative inexpensiveness, and potential for integration withstandard IC fabrication and packaging technology, both MEMS and BAWdevices are potentially attractive for applications in integrated RF LOclock generation. However, because of their manufacturing tolerances,both have significant part-to-part center frequency variation andadditionally have significant temperature variation which must be takeninto account. Finally, neither is tunable over the range of frequenciesrequired for a modern multi-carrier RF system.

Integer-N PLL technology is capable of producing low Phase Noise at itsoutput, provided it has either a low-Phase Noise Reference and a highPLL bandwidth or a low Phase Noise VCO and a low PLL bandwidth. To havea flexible output (controllable frequency output over a pre-determinedfrequency range), it requires either an extremely low frequency inputand a high multiplication factor or a flexible frequency input. Takentogether, these characteristics make it a poor fit on its own tocreating an integrated Low Phase Noise LO clock source for a modernmulti-carrier RF system.

Fractional-N PLL technology can provide extremely good frequencyflexibility and can operate from relatively high frequency referenceclocks, but normally requires relatively low PLL bandwidths in order toattenuate the Fractional-N Modulator noise from the output, which inturn requires either a high Postscaler divider ratio or a low PhaseNoise VCO in order to produce low Phase Noise output. The low PLLbandwidth is compatible with the reference clock characteristics,however the high Postscaler divider ratio is often incompatible with therequired LO frequency. As a result, a Fractional-N PLL taken on its ownis also a poor fit to creating an integrated Low Noise LO clock source.

Therefore, improvements in low noise LO clock signal sources aredesirable.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present disclosure will now be described, by way ofexample only, with reference to the attached Figures.

FIG. 1 illustrates a known Quadrature RF Receiver.

FIG. 2 illustrates a known Quadrature RF Transmitter.

FIG. 3A illustrates a known Single-Frequency Receive RF System FrequencyPlan prior to signals being mixed.

FIG. 3B illustrates the Single-Frequency Receive RF System FrequencyPlan of FIG. 3A after mixing the signals of FIG. 3A.

FIG. 4A illustrates a known Multi-Carrier Receive RF System FrequencyPlan prior to signals being mixed.

FIG. 4B illustrates the Multi-Carrier RF System Frequency Plan of FIG.4A after mixing the signals of FIG. 4A.

FIG. 5 illustrates a known Elementary Phase Locked Loop.

FIG. 6 illustrates a known Integer-N Phase Locked Loop.

FIG. 7 illustrates a known Fractional-N PLL.

FIG. 8 illustrates a Frequency Translation Mixer according to anembodiment of the present disclosure.

FIG. 9A illustrates a Frequency Translation Mixer Frequency Plan, withtwo pure tones and associated noise side-bands, prior to mixing thesignals and noise bands, according to an embodiment of the presentdisclosure.

FIG. 9B illustrates a sum term and a difference term resulting frommixing the pure tones and noise side bands of FIG. 9A.

FIG. 10 illustrates a 2-PLL Cascaded Up-Conversion Based LO ClockGenerator according to an embodiment of the present disclosure.

FIG. 11 illustrates a 2-PLL Nested Up-Conversion Based LO ClockGenerator according to an embodiment of the present disclosure.

FIG. 12 illustrates a 2-PLL Nested Down-Conversion Based LO ClockGenerator according to an embodiment of the present disclosure.

FIG. 13 illustrates a 3-PLL Nested Down-Conversion Based LO ClockGenerator according to an embodiment of the present disclosure.

FIG. 14A illustrates a Triple-Input Mixer Integer-N PLL according to anembodiment of the present disclosure.

FIG. 14B illustrates the triple-input mixer integer N PLL of FIG. 14Aintegrated in a Nested Down-Conversion Based LO Clock Generatoraccording to an embodiment of the present disclosure.

FIG. 14C illustrates the triple-input mixer integer N PLL of FIG. 14Aintegrated in a Nested Down-Conversion Based LO Clock Generatoraccording to an embodiment of the present disclosure.

FIG. 15 illustrates a 3-PLL Nested Down-Conversion Based LO ClockGenerator with Optimized Phase Detector Clocking according to anembodiment of the present disclosure.

FIG. 16 illustrates another 3-PLL Nested Down-Conversion Based LO ClockGenerator with Optimized Phase Detector Clocking according to anembodiment of the present disclosure.

FIG. 17 illustrates an embodiment of a method in accordance with thepresent disclosure.

FIG. 18 illustrates another embodiment of a method in accordance withthe present disclosure.

FIG. 19 illustrates yet another embodiment of a method in accordancewith the present disclosure.

DETAILED DESCRIPTION

The present disclosure provides a low Phase Noise LO clock generatorthat uses Integer-N and Fractional-N PLLs in combination with a mixerthat combines a high Phase Noise clock signal with a low Phase Noiseclock signal. The high Phase Noise clock signal can be a network clocksignal and the low Phase Noise clock signal can be that of, for example,a MEMS clock source or a BAW resonator clock source. The low Phase Noiseclock source (e.g., the MEMS clock source or the BAW resonator clocksource) can be integrated with the Integer-N and Fractional-N PLLs. TheLO clock generators in accordance with the present disclosure are highlyintegrated, relatively inexpensive, and produce low Phase Noise LOclocks suitable for use in modern multi-carrier RF systems.

Embodiments of LO clock generators described in the present disclosureprovide integrated low Phase Noise LO Clock generators for use in modernmulti-carrier RF systems. The LO clocks (clock signals) need to be ofsufficient quality for use in GSM applications, allow for multiplecarriers covering frequencies up to 100 MHz, cover applicationsrequiring LO clocks from approximate 700 MHz to 2 GHz, and providefrequency resolution of approximately 100 kHz.

In order to accomplish this, embodiments of the present disclosure usetwo reference clocks. The first, used as a frequency reference, hasrelatively high Phase Noise and can be provided by a recovered clockfrom a network but is frequency accurate. The second, used as a PhaseNoise reference, has very good Phase Noise and can be generated by, forexample, a high-Q local MEMS or BAW based resonator, but can haverelatively poor frequency accuracy due to temperature effects on theresonator and due to resonator manufacturing tolerances. In addition tothe two reference clocks, at least one Integer-N PLL is used to generatethe final output clock, a mixer is used to combine the two referenceclocks, and at least one Fractional-N PLL is added to allow for finetuning of the overall system.

In the context of the present disclosure, elements can be said to beoperationally connected to each other when, for example, a signalpresent in one element can be communicated to another element. Further,elements can be said to be operationally connected when an action in, orstate of, one element can be controlled by, or related to, an action in,or a state of, another element.

In a first aspect of the present disclosure, there is provided a clockgenerator unit to generate a target clock signal having a targetfrequency. The clock generator unit comprises: a Fractional-Nphase-locked loop (FNPLL) circuit having first phase detector circuitryand first voltage-controlled oscillator (VCO) circuitry operationallyconnected to the first phase detector circuitry, the first phasedetector circuitry having a FNPLL reference input terminal and a FNPLLfeedback input terminal, the FNPLL reference input terminal to receive afirst clock signal, the first clock signal having associated theretofirst clock signal Phase Noise, the FNPLL circuit being configured tohave a FNPLL bandwidth that filters out the first clock signal PhaseNoise and to generate an intermediate clock signal that is substantiallyfree of the first clock signal Phase Noise; an Integer-N PLL (INPLL)circuit having second phase detector circuitry and second VCO circuitryoperationally connected to the second phase detector circuitry, thesecond phase detector circuitry having an INPLL reference input terminaland an INPLL feedback input terminal, the INPLL reference input terminalto receive the intermediate clock signal, the second VCO circuitry tooutput the target clock signal, the INPLL circuit being configured tohave an INPLL bandwidth that filters out Phase Noise generated by thesecond VCO circuitry; mixer circuitry to output a mixer output signal;and filter circuitry to filter the mixer output signal. The mixercircuitry and the filter circuitry are formed in the FNPLL or in theINPLL. When the mixer circuitry and the filter circuitry are formed inthe FNPLL circuit, the mixer circuitry to receive a second clock signaland an output signal from the first VCO circuitry, the second clocksignal being unrelated to the first clock signal, the mixer circuitry togenerate a sum signal and a difference signal, the sum signal being at asum frequency that is a function of a sum of the frequency of the firstclock signal and the frequency of the output signal of the first VCOcircuitry, the difference signal being at a difference frequency that isa function a difference between the frequency of the first clock signaland the frequency of the output signal of the first VCO circuitry, thefilter circuitry to receive the sum signal and the difference signal andto generate a filtered signal that includes the sum signal, theintermediate signal being a function of the sum signal, the FNPLLfeedback input terminal configured to receive a FNPLL feedback signalthat is a function of a frequency and phase of the sum signal, the INPLLfeedback input terminal configured to receive an INPLL feedback signalthat is a function of the target signal. When the mixer circuitry andthe filter circuitry are formed in the INPLL circuit, the mixercircuitry to receive a second clock signal and an output signal from thesecond VCO circuitry, the second clock signal being unrelated to thefirst clock signal, the mixer circuitry to generate a sum signal and adifference signal, the sum signal being at a sum frequency that is afunction of a sum of the frequency of the first clock signal and thefrequency of the output signal of the second VCO circuitry, thedifference signal being at a difference frequency that is a function ofa difference between the frequency of the first clock signal and thefrequency of the output signal of the second VCO circuitry, the filtercircuitry to receive the sum signal and the difference signal and toproduce a filtered signal that includes the difference signal, the INPLLfeedback input terminal configured to receive an INPLL feedback signalthat is a function of the difference signal, the FNPLL feedback inputterminal configured to receive a feedback signal that is a function ofat least an output signal of the second VCO circuitry.

In a further aspect, the present disclosure provides clock generatorunit to generate a target clock signal having a target frequency. Theclock generator unit comprises: a first Fractional-N phase-locked loop(FFNPLL) circuit having first phase detector circuitry, firstvoltage-controlled oscillator (VCO) circuitry and first Fractional-Ndivider circuitry, the first phase detector circuitry beingoperationally connected to the first VCO circuitry, the first phasedetector circuitry having a FFNPLL reference input terminal and a FFNPLLfeedback input terminal, the FFNPLL reference input terminal to receivea first clock signal, the first clock signal having associated theretofirst clock signal Phase Noise, the FNPLL circuit being configured tohave a FNPLL bandwidth that filters out the first clock signal PhaseNoise, the FFNPLL to generate an intermediate clock signal in accordancewith the first clock signal and in accordance with a FFNPLL feedbacksignal received at the FFNPLL reference input terminal, the FFNPLLfeedback signal being a function of an output from the first VCOcircuitry, the intermediate clock signal being substantially free of thefirst clock signal Phase Noise; an Integer-N PLL (INPLL) circuit having:second phase detector circuitry; second VCO circuitry operationallyconnected to the second phase detector circuitry, the second phasedetector circuitry having an INPLL reference input terminal and an INPLLfeedback input terminal, the INPLL reference input terminal to receivethe intermediate clock signal; mixer circuitry; and filter circuitry,the mixer circuitry to receive a second clock signal and an outputsignal from the second VCO circuitry, the second clock signal beingunrelated to the first clock signal, the mixer circuitry to generate asum signal and a difference signal, the sum signal being at a sumfrequency that is a function of a sum of the frequency of the firstclock signal and the frequency of the output signal of the second VCOcircuitry, the difference signal being at a difference frequency that isa function of a difference between the frequency of the first clocksignal and the frequency of the output signal of the second VCOcircuitry, the filter circuitry to receive the sum signal and thedifference signal and to produce a filtered signal that includes thedifference signal, the INPLL feedback input terminal configured toreceive an INPLL feedback signal that is a function of the filteredsignal, the second VCO circuitry to output the target clock signal inaccordance with the intermediate signal and in accordance with the INPLLfeedback signal; and a second Fractional-N phase-locked loop (SFNPLL)circuit having third phase detector circuitry and second Fractional-Ndivider circuitry, the third phase detector circuitry having a SFNPLLreference input terminal and a SFNPLL feedback input terminal, theSFNPLL reference input terminal to receive the first clock signal, thesecond Fractional-N divider circuitry to receive the output signal fromthe second VCO circuitry and to generate an additional feedback signalas a function of settings of the SFNPLL, the SFNPLL feedback inputterminal to receive the additional feedback signal, the third phasedetector circuitry to output a control signal that is a function of thefirst clock signal and of the additional feedback signal, the firstFractional-N divider circuitry to receive the control signal, the FFNPLLfeedback signal also being a function of the control signal.

In a further aspect, the present disclosure provides a clock generatorunit to generate a target clock signal having a target frequency. Theclock generator unit comprises: a first Fractional-N phase-locked loop(FFNPLL) circuit having first phase detector circuitry, firstvoltage-controlled oscillator (VCO) circuitry and first Fractional-Ndivider circuitry, the first phase detector circuitry beingoperationally connected to the first VCO circuitry, the first phasedetector circuitry having a FFNPLL reference input terminal and a FFNPLLfeedback input terminal, the FFNPLL reference input terminal to receivea first clock signal, the first clock signal having associated theretofirst clock signal Phase Noise, the FNPLL circuit being configured tohave a FNPLL bandwidth that filters out the first clock signal PhaseNoise, the FFNPLL to generate an intermediate clock signal in accordancewith the first clock signal and in accordance with a FFNPLL feedbacksignal received at the FFNPLL reference input terminal, the FFNPLLfeedback signal being a function of an output from the first VCOcircuitry, the intermediate clock signal being substantially free of thefirst clock signal Phase Noise; an Integer-N PLL (INPLL) circuit having:second phase detector circuitry; second VCO circuitry operationallyconnected to the second phase detector circuitry, the second phasedetector circuitry having an INPLL reference input terminal and an INPLLfeedback input terminal, the INPLL reference input terminal to receivethe intermediate clock signal; mixer circuitry; and filter circuitry,the mixer circuitry to receive a second clock signal and an outputsignal from the second VCO circuitry, the second clock signal beingunrelated to the first clock signal, the mixer circuitry to generate asum signal and a difference signal, the sum signal being at a sumfrequency that is a function of a sum of the frequency of the firstclock signal and the frequency of the output signal of the second VCOcircuitry, the difference signal being at a difference frequency that isa function of a difference between the frequency of the first clocksignal and the frequency of the output signal of the second VCOcircuitry, the filter circuitry to receive the sum signal and thedifference signal and to produce a filtered signal that includes thedifference signal, the INPLL feedback input terminal configured toreceive an INPLL feedback signal that is a function of the filteredsignal, the second VCO circuitry to output the target clock signal inaccordance with the intermediate signal and in accordance with the INPLLfeedback signal; and a second Fractional-N phase-locked loop (SFNPLL)circuit having third phase detector circuitry and second Fractional-Ndivider circuitry, the third phase detector circuitry having a SFNPLLreference input terminal and a SFNPLL feedback input terminal, theSFNPLL reference input terminal to receive the FFNPLL feedback signal,the second Fractional-N divider circuitry to receive the output signalfrom the second VCO circuitry and to generate an additional feedbacksignal as a function of settings of the SFNPLL, the SFNPLL feedbackinput terminal to receive the additional feedback signal, the thirdphase detector circuitry to output a control signal that is a functionof the first clock signal and of the additional feedback signal, thefirst Fractional-N divider circuitry to receive the control signal, theFFNPLL feedback signal also being a function of the control signal.

In a further aspect, the present disclosure provides a clock generatorunit to generate a target clock signal having a target frequency. Theclock generator unit comprises: a Fractional-N phase-locked loop (FNPLL)circuit having phase detector circuitry and first voltage-controlledoscillator (VCO) circuitry operationally connected to the first phasedetector circuitry, the first phase detector circuitry having areference input terminal and a feedback input terminal, the referenceinput terminal to receive a first clock signal, the feedback inputterminal to receive a feedback signal, the first clock signal havingassociated thereto first clock signal Phase Noise, the FNPLL circuitbeing configured to have a FNPLL bandwidth that filters out the firstclock signal Phase Noise and to generate an intermediate clock signalthat is substantially free of the first clock signal Phase Noise, theintermediate clock signal being a function of the first clock signal andof the feedback signal; an Integer-N PLL (INPLL) circuit having atriple-input mixer circuitry, filter circuitry and second VCO circuitryoperationally connected to the mixer circuitry, the mixer circuitry toreceive the intermediate signal, a second clock signal and an outputsignal from the second VCO circuitry, the second clock signal beingunrelated to the first clock signal, the filter circuitry to receive anoutput from the mixer circuitry and to provide a filtered output signalto the second VCO, the second VCO circuitry to output the target clocksignal, the feedback signal being a function of the target clock signal.

In a further aspect, the present disclosure provides a clock generatorunit to generate a target clock signal having a target frequency, theclock generator unit comprises: a first Fractional-N phase-locked loop(FFNPLL) circuit having first phase detector circuitry, firstvoltage-controlled oscillator (VCO) circuitry and first Fractional-Ndivider circuitry, the first phase detector circuitry beingoperationally connected to the first VCO circuitry, the first phasedetector circuitry having a FFNPLL reference input terminal and a FFNPLLfeedback input terminal, the FFNPLL reference input terminal to receivea first clock signal, the first clock signal having associated theretofirst clock signal Phase Noise, the FNPLL circuit being configured tohave a FNPLL bandwidth that filters out the first clock signal PhaseNoise, the FFNPLL to generate an intermediate clock signal in accordancewith the first clock signal and in accordance with a FFNPLL feedbacksignal received at the FFNPLL feedback input terminal, the FFNPLLfeedback signal being a function of an output from the first VCOcircuitry, the intermediate clock signal being substantially free of thefirst clock signal Phase Noise; an Integer-N PLL (INPLL) circuit havinga triple-input mixer circuitry, filter circuitry and second VCOcircuitry operationally connected to the mixer circuitry, the mixercircuitry to receive the intermediate signal, a second clock signal andan output signal from the second VCO circuitry, the second clock signalbeing unrelated to the first clock signal, the filter circuitry toreceive an output from the mixer circuitry and to provide a filteredoutput signal to the second VCO, the second VCO circuitry to output thetarget clock signal; and a second Fractional-N phase-locked loop(SFNPLL) circuit having second phase detector circuitry and secondFractional-N divider circuitry, the third phase detector circuitryhaving a SFNPLL reference input terminal and a SFNPLL feedback inputterminal, the SFNPLL reference input terminal to receive the first clocksignal, the second Fractional-N divider circuitry to receive the outputsignal from the second VCO circuitry and to generate an additionalfeedback signal as a function of settings of the SFNPLL, the SFNPLLfeedback input terminal to receive the additional feedback signal, thesecond phase detector circuitry to output a control signal that is afunction of the first clock signal and of the additional feedbacksignal, the first Fractional-N divider circuitry to receive the controlsignal, the FFNPLL feedback signal also being a function of the controlsignal.

In yet another aspect, the present disclosure provides a clock generatorunit to generate a target clock signal having a target frequency, theclock generator unit comprises: a first Fractional-N phase-locked loop(FFNPLL) circuit having first phase detector circuitry, firstvoltage-controlled oscillator (VCO) circuitry and first Fractional-Ndivider circuitry, the first phase detector circuitry beingoperationally connected to the first VCO circuitry, the first phasedetector circuitry having a FFNPLL reference input terminal and a FFNPLLfeedback input terminal, the FFNPLL reference input terminal to receivea first clock signal, the first clock signal having associated theretofirst clock signal Phase Noise, the FNPLL circuit being configured tohave a FNPLL bandwidth that filters out the first clock signal PhaseNoise, the FFNPLL to generate an intermediate clock signal in accordancewith the first clock signal and in accordance with a FFNPLL feedbacksignal received at the FFNPLL reference input terminal, the FFNPLLfeedback signal being a function of an output from the first VCOcircuitry, the intermediate clock signal being substantially free of thefirst clock signal Phase Noise; an Integer-N PLL (INPLL) circuit havinga triple-input mixer circuitry, filter circuitry and second VCOcircuitry operationally connected to the mixer circuitry, the mixercircuitry to receive the intermediate signal, a second clock signal andan output signal from the second VCO circuitry, the second clock signalbeing unrelated to the first clock signal, the filter circuitry toreceive an output from the mixer circuitry and to provide a filteredoutput signal to the second VCO, the second VCO circuitry to output thetarget clock signal; and an second Fractional-N phase-locked loop(SFNPLL) circuit having third phase detector circuitry and secondFractional-N divider circuitry, the third phase detector circuitryhaving a SFNPLL reference input terminal and a SFNPLL feedback inputterminal, the SFNPLL reference input terminal to receive the FFNPLLfeedback signal, the second Fractional-N divider circuitry to receivethe output signal from the second VCO circuitry and to generate anadditional feedback signal as a function of settings of the SFNPLL, theSFNPLL feedback input terminal to receive the additional feedbacksignal, the third phase detector circuitry to output a control signalthat is a function of the first clock signal and of the additionalfeedback signal, the first Fractional-N divider circuitry to receive thecontrol signal, the FFNPLL feedback signal also being a function of thecontrol signal.

In yet another aspect, the present disclosure provides a method togenerate a target clock signal. The method comprises: at a Fractional-Nphase locked loop (FNPLL) circuit: receiving a first clock signal;receiving a second clock signal, the second clock signal being unrelatedto the first clock signal; generating a modified clock signal as afunction of the first clock signal and as a function of the second clocksignal; with a mixer, combining the modified clock signal with thesecond clock signal to obtain a signal having a frequency sum componentand a frequency difference component; and outputting an intermediateclock signal having a frequency that is a function of the frequency ofthe frequency sum component, the FNPLL having an FNPLL bandwidthselected to filter out Phase Noise present in the first clock signal.The method further comprises at Integer-N phase locked loop (INPLL)circuit: receiving the intermediate clock signal; and outputting thetarget clock signal as a function of the intermediate clock signal andas a function of a feedback signal, the INPLL circuit having an INPLLbandwidth selected to filter out Phase Noise generated by components ofthe INPLL circuit.

In another aspect, the present disclosure provides a method to generatea target clock signal, the method comprising: at a Fractional-N phaselocked loop (FNPLL) circuit:

receiving a first clock signal; and generating an intermediate clocksignal as a function of the first clock signal and as a function of afirst feedback signal, the FNPLL having an FNPLL bandwidth selected tofilter out Phase Noise present in the first clock signal. The methodfurther comprises, at an Integer-N phase locked loop (INPLL) circuit:receiving the intermediate clock signal; receiving a second clocksignal, the second clock signal being unrelated to the first clocksignal; with a mixer, combining the intermediate clock signal with thesecond clock signal to obtain a signal having a frequency sum componentand a frequency difference component; and generating a second feedbackclock signal that is a function of the frequency difference component;outputting the target clock signal as a function of the intermediateclock signal and as a function of the second feedback signal, the INPLLcircuit having an INPLL bandwidth selected to filter out Phase Noisegenerated by components of the INPLL circuit.

In yet another aspect, the present disclosure provides a method togenerate a target clock signal. The method comprises: at a Fractional-Nphase locked loop (FNPLL) circuit: receiving a first clock signal; andgenerating an intermediate clock signal as a function of the first clocksignal and as a function of a first feedback signal, the FNPLL having anFNPLL bandwidth selected to filter out Phase Noise present in the firstclock signal. The method further comprises, at an Integer-N phase lockedloop (INPLL) circuit: receiving the intermediate clock signal; receivinga second clock signal, the second clock signal being unrelated to thefirst clock signal; with a triple-input mixer, combining theintermediate clock signal with the second clock signal and with a secondfeedback signal received from voltage-controlled oscillator (VCO)circuitry of the INPLL circuit to obtain a signal having signalcomponents at a plurality of frequencies, one of the plurality offrequencies being the closest to DC, one of the resulting signalcomponents being at the frequency closest to DC; controlling the VCOcircuitry as a function of the resulting signal at the frequency closestto DC; and outputting the target clock signal, the target signal being afunction of the second feedback signal, the INPLL circuit having anINPLL bandwidth selected to filter out Phase Noise generated bycomponents of the VCO circuitry.

Mixers as Frequency Translation Elements

At the core of embodiments of the present disclosure is the notion touse RF mixers to translate one frequency to another without dividers,thereby avoiding Phase Noise multiplication that would otherwise happenin a classic PLL. This approach is shown in FIG. 8, which shows an RFmixer 801 that combines two clocks, one a Frequency Reference and theother a Phase Noise Reference. The output from the mixer is fed througha filter 802 to remove unwanted images.

The frequency plan for such a mixer is shown in FIGS. 9A and 9B. FIG. 9Ashows that the inputs to the mixer 801 are two pure tones 901 and 911,with their associated Phase Noise side-bands 902 and 912. As establishedby Equation 2 above and as shown at FIG. 9B, the output from the mixer801 includes the sum term 921, the difference term 931, and their PhaseNoise side-bands 922 and 932 respectively.

In general, outputs from the mixer will actually be the convolution (inthe frequency domain) of the two input signals (including Phase Noiseside-bands), however for practical clock signal power levels, the outputPhase Noise side-bands 922 and 932 power levels can be taken as the sumof the input Phase Noise side bands 902 and 912 power levels, whichsimplifies analysis.

Using RF mixers for frequency translation is not a new concept, howeverprior art in this area has focused on fine frequency synthesis, using amixer to combine a high-frequency coarsely-tuned clock signal with alow-frequency finely-spaced clock signal to create a high-frequencyfinely-spaced clock signal as an alternative to using a Fractional-NPLL. Embodiments of the present disclosure, however, focus on the PhaseNoise implications of mixing, not on fine frequency tuning; aFractional-N PLL provides that functionality. Mathematically, inaccordance with the present disclosure, the mixer provides an additionor subtraction operation, which allows LO clock generator designers tochoose circuit topologies to minimize Phase Noise gains from noisyfrequency references.

With reference to FIGS. 8, 9A and 9B, for applications where the mixeroperates as an Up-Conversion mixer, and the desired frequency output isthe sum term 921, the filter 802 is a band-pass filter that passes thesum term and rejects the difference term 931. In this application, theFrequency Reference is a comparatively low frequency input compared tothe Phase Noise Reference.

For applications where the mixer operates as a Down-Conversion mixer,and the desired frequency output is the difference term 931, the filter802 is a low-pass filter that passes the difference term and rejects thesum term 921. In this application, the Frequency Reference is at acomparable frequency to the Phase Noise Reference, and is generated by aPLL with a comparatively wide bandwidth to attenuate Phase Noisegenerated by the PLL components.

As an example of an Up-Conversion mixer application, consider asituation where the LO system designer has a 100 MHz FrequencyReference, and requires a 2700 MHz Output Clock frequency. In aconventional Integer-N PLL (FIG. 6) the three dividers would beconfigured to provide a multiply-by-27 frequency operation from theFrequency Reference to the Output Clock (Equation 3). As a result thePhase Noise in the PLL bandwidth will also be multiplied by 27× (orshifted up by 20×log₁₀(27)≈29 dBc/Hz). In order to limit the amount ofPhase Noise present at the output due to the Frequency Reference, thePLL bandwidth must be made low, however in that case the Phase Noisefrom the VCO will dominate.

If the LO generator designer also has a 2600 MHz Phase Noise Reference(for example, a low Phase Noise BAW-based reference) and a mixer, he/shecan mix the 100 MHz Frequency Reference with the 2600 MHz Phase Noise,producing 2500 MHz (the “difference”) and 2700 MHz (the “sum”) outputs.Using a filter to reject the 2500 MHz output leaves the desired 2700 MHzoutput. The frequency (and therefore phase) gain from the input to theoutput is 1, for a 27× reduction in Phase Noise at the output due to theFrequency Reference. However, the 2600 MHz Phase Noise reference willhave frequency inaccuracies due to manufacturing tolerances andtemperature effects and therefore, the mixer and filter will need to beembedded in a PLL to accurately track the Frequency Reference. Inaddition, the desired output frequency will not be as simple to produceas in this example, and a PLL (often a Fractional-N PLL) will berequired in order to provide the required frequency.

A First Embodiment of an Up-Conversion Based LO Clock Generator:

A first exemplary embodiment of an Up-Conversion based LO clockgenerator is shown in FIG. 10. The Up-Conversion based LO clockgenerator comprises a Fractional-N PLL 1000 cascaded with an Integer-NPLL 1010. The fractional-N PLL 1000 receives a frequency referencesignal 5 and outputs an intermediate signal 10. The Fractional-N PLLdiffers from the one described in FIG. 7 in that a Mixer 1008 isintroduced into the output path that mixes the output from the VCO 1003with a low Phase Noise Reference signal 15 sourced from a Phase NoiseReference Source (generator) 500 (e.g., a MEMS resonator clock source, aBAW resonator clock source, a surface acoustic wave resonator clocksource, a quartz clock source, etc.) The mixer 1008 operates in anUp-Conversion mode, and a band-pass filter 1009 (filter) is used toselect the “sum” output from the mixer. The Fractional-N PLL 1000 isconfigured to produce a convenient reference frequency (of theintermediate signal 10) for the Integer-N PLL 1010, which in turnproduces the desired output frequency (the output clock 20). Theintermediate clock signal is substantially free of the first clocksignal Phase Noise, which is to be understood as meaning at least thatthe signal to noise ratio of the intermediate clock signal is betterthan that of the first clock signal.

An additional divider 1006 (frequency divider) in the Fractional-N PLL1000 is configured to produce a relatively large divide ratio, whichacts to attenuate Phase Noise generated by the VCO 1003. A lowFractional-N PLL bandwidth attenuates Phase Noise from the FrequencyReference and from the Fractional-N divider 1004 and 1005, while a wideInteger-N PLL bandwidth will attenuate Phase Noise that comes from theInteger-N PLL VCO component.

The Fractional-N PLL 1000 can also be referred to as a fractional-N PLLcircuit and the Integer-N PLL 1010 can also be referred to as anInteger-N PLL circuit. The phase detectors 1001 and 1011 can each bereferred to as phase detector circuitry that may include a loop filter1002 and 1012 respectively as well as a prescaler divider 1007 and 1017respectively. Each of the phase detectors (each of the phase detectorcircuitries) has a reference input terminal and a feedback inputterminal. The VCOs 1003 and 1013 can each be referred to as VCOcircuitry that may include a postscaler divider (frequency divider) 1006or 1016. The fractional-N divider 1005 and 1005 can be referred to asfractional-N divider circuitry. The filter (or band pass filter) 1009can be referred to as filter circuitry. The phase detector 1001 receivesa feedback signal that is a function of the sum signal generated at theoutput of the band bass filter 1009. In the present embodiment, thefeedback signal received at the feedback input terminal of the phasedetector 1001 is at a frequency that is equal to a rational fraction ofthe frequency of the sum signal. The feedback signal received at thefeedback input terminal of the phase detector 1011 is a function of thetarget clock signal (output clock) output from the postscaler divider1016. In the present embodiment, the feedback signal received at thephase detector 1011 is at a frequency equal to that of the target clocksignal multiplied by the inverse of divide ratio of postscaler divider1016, divided by the divide ratio of the divider 1014.

A Second Up-Conversion Based LO Clock Generator:

A second LO clock generator exemplary embodiment of the presentdisclosure is shown in FIG. 11. Similar to the generator shown in FIG.10, the embodiment of FIG. 11 comprises a Fractional-N PLL 1100, a PhaseNoise reference source 500 and an Integer-N PLL 1010, however in thisconfiguration the two PLLs are nested, not cascaded. FIG. 11 also showsa frequency reference signal 5, an intermediate signal 10, a Phase Noisereference signal 15 and an output clock 20.

Nesting the PLLs, that is, embedding the Integer-N PLL (1010) inside theFractional-N PLL, has similar properties to cascading them, with themajor exception that the Fractional-N divider (1104/1105) is configureddifferently, taking its input from the VCO 1013 of the Integer-N PLL1010 rather than from the filter or band pass filter 1009. Followingclassic control theory on nested feedback loops, so long as theInteger-N PLL's bandwidth is much greater than the Fractional-N PLL'sbandwidth, the loop will be stable. The major advantage of thisconfiguration over the cascaded alternative is that the final frequencyselection is somewhat simplified because the output clock frequency isnow dependent on 3 dividers, rather than 5.

Both LO Clock Generators of FIG. 10 and FIG. 11 provide a flexible LOclock out through the use of a Fractional-N divider. The Phase Noise ofthe Frequency Reference is passed through to the output with amultiplication factor set by the feedback dividers, filtered by the PLLbandwidth of the Fractional-N PLL. The VCO noise of the Fractional-N PLLis attenuated by a large divider above the Fractional-N PLL's bandwidth,and is summed with the low Phase Noise provided by the Phase NoiseReference. This sum is a low Phase Noise Reference for the Integer-NPLL, which has a wide PLL bandwidth to suppress its internal noisesources, and as a result the overall Phase Noise at the Output Clockremains relatively low noise.

The band-pass filter 1009 of the Up-Conversion based LO Clock Generatorsof FIG. 10 and FIG. 11 can be difficult to design. This is because inorder to attenuate Phase Noise from the VCO 1003 the divider 1006 mustbe made relatively large, and therefore the frequency input to the mixer1008 from the divider 1006 is relatively small, and the frequencyspacing between the “sum” and “difference” outputs is also relativelysmall. This in turn requires that the band-pass filter 1009 have arelatively narrow pass-band, which in turn requires that the filter havea relatively high order and in turn requires either tight componenttolerances or a method of tuning the filter against loose componenttolerances. In addition, if the LO Clock Generator is required to beflexible and cover a wide range of frequencies, the filter must also betunable to cover the extended frequency range.

The Fractional-N PLL 1100 can also be referred to as a fractional-N PLLcircuit and the Integer-N PLL 1010 can also be referred to as anInteger-N PLL circuit. The phase detectors 1001 and 1011 can each bereferred to as phase detector circuitry that may include a loop filter1002 and 1012 respectively as well as a prescaler divider 1007 and 1017respectively. Each of the phase detectors (each of the phase detectorcircuitries) has a reference input terminal and a feedback inputterminal. The VCOs 1003 and 1013 can each be referred to as VCOcircuitry that may include a postscaler divider (frequency divider) 1006or 1016. The fractional-N divider 1005 and 1005 can be referred to asfractional-N divider circuitry. The filter (or band pass filter) 1009can be referred to as filter circuitry. The phase detector 1001 receivesa feedback signal that is a function of the sum signal generated at theoutput of the band bass filter 1009. In the present embodiment, thefeedback signal received at the feedback input terminal of the phasedetector 1001 is at a frequency that is equal to a rational fraction ofthe frequency of the sum signal. The feedback signal received at thefeedback input terminal of the phase detector 1001 is a function of thetarget clock signal (output clock) output from the postscaler divider1016. It is in fact a function of the frequency and phase of the targetclock signal. In the present embodiment, the feedback signal received atthe phase detector 1001 is at a frequency equal to that of the targetclock signal multiplied by the inverse of divide ratio of postscalerdivider 1016, divided by the divide ratio of the divider 1104. As thetarget clock signal depends from the sum signal provided by thefractional-N PLL 1100, the feedback signal provided to the phasedetector 1001 is also a function of the sum signal. The feedback signalreceived at the phase detector 1001 is at a frequency equal to that ofthe target clock signal multiplied by the inverse of divide ratio ofpostscaler divider 1016, divided by the divide ratio of the divider1014.

A First Down-Conversion Based LO Clock Generator:

An alternative embodiment, one that avoids the above-noted issue withthe band-pass filter 1009 involves a Down-Conversion mixer. An LO clockgenerator that implements this approach is shown in FIG. 12. As in FIG.10 and FIG. 11, the generator uses a Fractional-N PLL 1200, an Integer-NPLL 1210 and a Phase Noise reference source 500, and as in FIG. 11 theInteger-N PLL is nested inside the Fractional-N PLL. However in thisconfiguration, the Phase Noise Reference and Mixer 1218 are insertedinto the feedback path of the Integer-N PLL, and the filter 1219 thatfollows the mixer is a low-pass filter rather than a band-pass filter.FIG. 12 also shows a frequency reference signal 5, an intermediatesignal 10, a Phase Noise reference signal 15 and an output clock 20.

As in FIG. 10 and FIG. 11 the output from the Fractional-N PLL VCO 1203is divided by a large-value divider 1206, and then the Phase NoiseReference frequency is added, however because the mixer 1218 is placedin the Integer-N feedback path and is used as a Down-Conversion mixer,this addition is performed by subtracting the Phase Noise Referencefrequency from a similar-frequency output from feedback divider 1214resulting in a low frequency, rather than by addition of a low frequencyto a high frequency as in the Up-Conversion designs.

Mathematically, the Output Clock from FIG. 12 is almost identical tothat of FIG. 11, however the Down-Conversion operation (subtracting twohigh frequencies to produce a low frequency instead of adding a lowfrequency to a high frequency) means that there is a much widerfrequency spacing between the sum and difference outputs from the mixer,which greatly simplifies the filter design, and furthermore allow the LOClock Generator to use a simpler Low-Pass Filter instead of a Band-PassFilter.

As in other embodiments, the Fractional-N PLL 1200 can also be referredto as a fractional-N PLL circuit and the Integer-N PLL 1210 can also bereferred to as an Integer-N PLL circuit. The phase detectors 1201 and1211 can each be referred to as phase detector circuitry that mayinclude a loop filter 1202 and 1212 respectively as well as a prescalerdivider such as, for example the prescaler divider 1207. Each of thephase detectors (each of the phase detector circuitries) has a referenceinput terminal and a feedback input terminal. The VCOs 1203 and 1213 caneach be referred to as VCO circuitry that may include a postscalerdivider (frequency divider) 1206 or 1216. The fractional-N divider 1205and 1204 can also be referred to as fractional-N divider circuitry. Thefilter (or low pass filter) 1219 can be referred to as filter circuitry.The phase detector 1201 receives a feedback signal that is a function ofthe output clock (target clock signal). In the present embodiment, thefeedback signal received at the feedback input terminal of the phasedetector 1201 is at a frequency that is equal to a rational fraction ofthe frequency of the sum signal. The feedback signal received at thefeedback input terminal of the phase detector 1201 is a function of thetarget clock signal (output clock) output from the VCO 1213. It is infact a function of the frequency and phase of the target clock signal.In the present embodiment, the feedback signal received at the phasedetector 1201 is at a frequency equal to that of the target clock signalmultiplied by the inverse of divide ratio of postscaler divider 1216,divided by the divide ratio of the divider 1204. As the target clocksignal depends from the intermediate signal provided by the fractional-NPLL 1200 to the Integer-N PLL 1210, the feedback signal provided to thephase detector 1201 is also a function of the intermediate signal. Thefeedback signal received at the phase detector 1201 is at a frequencyequal to that of the target clock signal multiplied by the inverse ofdivide ratio of postscaler divider 1216, divided by the divide ratio ofthe divider 1204.

A Second Down-Conversion Based LO Clock Generator:

Another Down-Conversion mixer based LO clock generator embodiment, inaccordance with the present disclosure is shown in FIG. 13 and comprisesa Phase Noise reference source 500, a Fractional-N PLL 1300 cascadedwith an Integer-N PLL 1210, plus a second Fractional-N PLL 1320 (i.e., athird PLL) with the other two PLLs nested inside of it. This secondFractional-N PLL 1320 controls the first fractional-N PLL 1300 bydriving the Fractional-N modulator 1305 using techniques similar tothose described in U.S. Pat. No. 7,986,190 B1, and is therefore afully-digital PLL with a fully-integrated digital loop filter 1322. FIG.13 also shows a frequency reference signal 5, an intermediate signal 10,a Phase Noise reference signal 15 and an output clock 20.

The inclusion of this third PLL (1320) allows the system to have a muchlower overall bandwidth than would otherwise be reasonable in anintegrated system, and allows for a greater degree of modularity inchoosing the various loop bandwidths: the first Fractional-N PLL 1300can have a bandwidth chosen to optimize the output Phase Noiseindependent of the required bandwidth to filter the Phase Noise of theFrequency Reference, provided that its bandwidth is greater than thebandwidth of the second Fractional-N PLL 1320 and less than thebandwidth of the Integer-N PLL 1310. The second Fractional-N PLL 1320with the digital loop filter 1322 may provide an extremely low loopbandwidth, rejecting the maximum amount of Phase Noise from theFrequency Reference as possible. Outside of the bandwidth of the secondfractional-N PLL 1320, the embodiment of FIG. 13 performs identically tothat in FIG. 12.

As in other embodiments, the Fractional-N PLL 1300 can also be referredto as a fractional-N PLL circuit and the Integer-N PLL 1210 can also bereferred to as an Integer-N PLL circuit. The phase detectors 1301 and1211 can each be referred to as phase detector circuitry that mayinclude a loop filter 1302 and 1212 respectively as well as a prescalerdivider such as, for example the prescaler divider 1307. Each of thephase detectors (each of the phase detector circuitries) has a referenceinput terminal and a feedback input terminal. The VCOs 1303 and 1213 caneach be referred to as VCO circuitry that may include a postscalerdivider (frequency divider) 1206 and 1216 respectively. The fractional-Ndivider 1305 and 1304 can also be referred to as fractional-N dividercircuitry. The filter (or low pass filter) 1219 can be referred to asfilter circuitry. The phase detector 1301 receives a feedback signalthat is a function of the intermediate signal provided by thefractional-N PLL 1300 to the Integer-N PLL 1210. In the presentembodiment, the feedback signal received at the feedback input terminalof the phase detector 1301 is at a frequency that is equal to that ofthe intermediate signal divided by the divide ratio of the frequencydivider 1304. The feedback signal received at the feedback inputterminal of the phase detector 1201 is a function of the target clocksignal (output clock) output from the VCO 1213, and, it is also afunction of the difference signal output from the low pass filter 1219.

Additional Embodiments and Design Optimizations:

There are many optimizations available in implementing embodiments ofthe present disclosure. One such optimization involves the Integer-N PLL1210, of FIG. 12 and FIG. 13, and provides a simplified circuit shown inFIG. 14A. This simplified Integer-N PLL 1410 can be substituted in placeof the Integer-N PLL 1210 of FIG. 12 and FIG. 13, and takes advantage ofthe fact that the PLL itself acts as a low-pass filter and the Low-PassFilter 1219 of FIGS. 12 and 13 can be removed, and of the fact that thePhase Detector 1211 can be implemented with a mixer, meaning that boththe original mixer 1218 and phase detector 1211 can be replaced with atriple-input mixer 1418, and the Low Pass filter 1219 can be removed.The Triple-Input-Mixer 1418 of FIG. 14A receives a Phase Noise referencesignal 2000, a feedback input signal 2002 from the VCO 1413, through afrequency divider 1414, and an input signal 2004 from a fractional-N PLLcircuit (not shown). The output from the Triple-Input-Mixer 1418 willhave frequency components at each possible combination (both sum anddifference) of the three frequencies of signals 2000, 2002 and 2004. Onesuch frequency component will be equal to the sum of the frequency ofthe Phase Noise reference signal 2000 and the frequency of the inputsignal 2004 provided by the Fractional-N PLL, minus the frequency of thefeedback input signal 2002. This frequency component will be at DC (0Hz) when the Integer-N PLL 1410 is locked or, if not locked, will bedriven to DC by the Integer-N PLL 1410. The low-pass response of thePLL, which includes the loop filter 1412, will tend to pass this desirednear-DC frequency component, and will tend to attenuate all otherundesirable frequency components.

FIG. 14B shows how the embodiment of FIG. 12 can be modified inaccordance with the Triple-Input Mixer Integer-N PLL of FIG. 14A. FIG.14C shows how the embodiment of FIG. 13 can be modified in accordancewith the Triple-Input Mixer Integer-N PLL of FIG. 14A. FIGS. 14B and 14Calso show a frequency reference signal 5, an intermediate signal 10, aPhase Noise reference signal 15 and an output clock 20.

One limitation of this optimization is that a mixer-based Phase Detectoris only capable of tracking frequencies over a limited range, and as aresult will likely need to use a second phase detector (not shown), suchas a Phase/Frequency Detector (PFD) during the initial lock.

Another optimization simplifies the clocking of the Phase Detector 1321associated with the second Fractional-N PLL 1320 in FIG. 13, and isshown in FIG. 15. This optimization modifies this second Fractional-NPLL 1520 (FIG. 15), driving the reference input to the phase detectorwith the feedback clock from the first PLL rather than from theFrequency Reference.

Within the bandwidth of the first Fractional-N PLL 1300, this feedbackclock tracks the Phase Noise of the Frequency Reference and, given thebandwidth of the first Fractional-N PLL, is greater than that of thesecond Fractional-N PLL 1520. The results are mathematically equivalent.This modification, however, significantly simplifies implementation ofthe system. Placing the Phase Detector 1521, Loop Filter 1422, andFractional-N Modulator 1305 on the same clock domain, simplifies datahand-off from one block to the next. In an example embodiment, apreferred implementation of the Phase Detector 1521 uses aTime-to-Digital Converter (TDC) that is compatible with standard digitalSynthesis and Place and Route implementation techniques that can be laidout together with the other parts of the system, thereby reducing theimplementation cost substantially compared to the original design. FIG.15 also shows a frequency reference signal 5, an intermediate signal 10,a Phase Noise reference signal 15 and an output clock 20.

FIG. 16 shows a variant of the embodiment of FIG. 15. The variant ofFIG. 16 comprises, rather than the Integer-N PLL 1210 of FIG. 15, anInteger-N PLL 1410 that comprises a triple-input mixer 1418, which canbe the same as used in the example of FIG. 14C.

FIG. 17 is a flowchart of a method according to the present disclosure.The method shown can be implemented using, for example, the embodimentof FIG. 10 or the embodiment of FIG. 11.

The method begins at a Fractional-N PLL (FNPLL) circuit where, at action1700, a first clock signal is received. The first clock signal can beFrequency Reference signal. Subsequently, at action 1702, a second clocksignal is received. The second clock signal is unrelated to the firstclock signal. “Unrelated” is to be understood as meaning that the secondclock signal is generated by clock source that is different from theclock source that generates the first clock signal. The second clocksignal can be a Phase Noise reference signal. At action 1704, a modifiedclock signal is generated as a function of the first clock signal and asa function of the second clock signal. Subsequently, at a mixercomprised in the Fractional-N PLL circuit, the modified clock signal iscombined, at action 1706, with the second clock signal to obtain asignal having a frequency sum component and a frequency differencecomponent. Following this, an intermediate clock that is a function ofthe frequency of the frequency sum component is output. The Fractional-NPLL circuit has an FNPLL bandwidth selected (engineered, designed) tofilter out Phase Noise present in the first clock signal.

At an Integer-N PLL circuit, the intermediate clock signal is receivedat action 1710 and, at action 1712, a target clock signal is output as afunction of the intermediate clock signal and as a function of afeedback signal. The Integer-N PLL circuit has an INPLL bandwidthselected (engineered, designed) to filter out Phase Noise generated bycomponents of the INPLL circuit.

FIG. 18 is a flowchart of a method according to the present disclosure.The method shown can be implemented using, for example, the embodimentof FIG. 12, Figure or FIG. 15.

The method begins at a Fractional-N PLL circuit where, at action 1800, afirst clock signal is received. The first clock signal can be FrequencyReference signal. At action 1802, an intermediate clock signal isgenerated as a function of the first clock signal and as a function of afirst feedback signal. The Fractional-N PLL has a bandwidth selected tofilter out Phase Noise present in the first clock signal.

Subsequently, at an Integer-N phase locked loop (INPLL) circuit, theintermediate clock signal is received at action 1804. At action 1806, asecond clock signal is received. The second clock signal is unrelated tothe first clock signal. Then, at action 1808, with a mixer, theintermediate clock signal is combined with the second clock signal toobtain a signal having a frequency sum component and a frequencydifference component. At action 1810, a second feedback clock signal isgenerated as a function of the frequency difference component. Finally,at action 1812, the target clock signal is output, the target clocksignal is a function of the intermediate clock signal and of the secondfeedback signal.

FIG. 19 is a flowchart of a method according to the present disclosure.The method shown can be implemented using, for example, the embodimentof FIG. 14B, FIG. 14C or FIG. 16.

The method begins at a Fractional-N PLL circuit where, at action 1900, afirst clock signal is received. The first clock signal can be FrequencyReference signal. At action 1902, an intermediate clock signal isgenerated as a function of the first clock signal and as a function of afirst feedback signal. The Fractional-N PLL has an FNPLL bandwidthselected to filter out Phase Noise present in the first clock signal.

Subsequently, at an Integer-N PLL circuit, the intermediate clock signalis received at action 1904 and, a second clock signal is received ataction 1906. The second clock signal is unrelated to the first clocksignal. Then, at action 1908, at a triple-input mixer comprised in theInteger-N PLL circuit, the intermediate clock signal is combined withthe second clock signal and with a second feedback signal received fromvoltage-controlled oscillator (VCO) circuitry of the Integer-N PLLcircuit. The output of the triple-input mixer is a resulting signal thathas signal components at a plurality of frequencies. One of theplurality of frequencies is the closest to DC and one of the resultingsignal components is at the frequency closest to DC. Subsequently, ataction 1910, VCO circuitry of the Integer-N PLL is controlled as afunction of the resulting signal component at the frequency closest toDC. Finally, at action 1912, the target clock signal is output, thetarget signal being a function of the second feedback signal. TheInteger-N PLL circuit has an INPLL bandwidth selected to filter outPhase Noise generated by components of its VCO circuitry.

There are many advantages that embodiments of the present disclosureallow in the construction of RF systems, including at least one of thefollowing: the creation of an integrated low Phase Noise LO clock thatmeets challenging RF Phase Noise requirements such as those in GSM;sufficient tuning range for multiple carrier applications coveringapplications requiring LO clocks from approximately 700 MHz to 2 GHz,while providing frequency resolution of approximately 100 kHz; and thelow Phase Noise LO clock generation is fully integrated with the rest ofthe RF sub-components in a single-package, single-die device. The PhaseNoise Reference can be based on, for example, a BAW or MEMS resonatormounted in an IC device package, or micro-machined on semiconductor die.

The embodiments of the present disclosure allow single-packageintegration of multiple components to create a high quality LO clocksource using a CMOS process. This application has traditionally requireda bipolar process, and often has required multiple devices in multiplepackages connected on a printed circuit board.

Generating a fully-integrated wide tuning range, low Phase Noise clockthat meets GSM RF specifications is a challenge and problem to solve. Anumber of methods have been disclosed to produce this wide tuning range,low Phase Noise clock for use in integrated Radio Frequency systems.

The methods all use two reference clocks: a frequency-accurate referencethat has comparatively high Phase Noise, and a frequency-inaccuratereference that has comparatively low Phase Noise. By combining multiplePhase-Locked Loops and at least one mixer, it has been shown howflexible output frequencies, whose frequency accuracy is derived fromthe first reference clock but whose Phase Noise is derived from thesecond reference clock, can be produced in a readily-integrated andrelatively low-cost system. Two general embodiments of the inventionhave been disclosed: Up-Conversion based and Down-Conversion based.

In the preceding description, for purposes of explanation, numerousdetails are set forth in order to provide a thorough understanding ofthe embodiments. However, it will be apparent to one skilled in the artthat these specific details are not required. In other instances,well-known electrical structures and circuits are shown in block diagramform in order not to obscure the understanding. For example, specificdetails are not provided as to whether the embodiments described hereinare implemented as a software routine, hardware circuit, firmware, or acombination thereof.

Embodiments of the disclosure can be represented as a computer programproduct stored in a machine-readable medium (also referred to as acomputer-readable medium, a processor-readable medium, or a computerusable medium having a computer-readable program code embodied therein).The machine-readable medium can be any suitable tangible, non-transitorymedium, including magnetic, optical, or electrical storage mediumincluding a diskette, compact disk read only memory (CD-ROM), memorydevice (volatile or non-volatile), or similar storage mechanism. Themachine-readable medium can contain various sets of instructions, codesequences, configuration information, or other data, which, whenexecuted, cause a processor to perform steps in a method according to anembodiment of the disclosure. Those of ordinary skill in the art willappreciate that other instructions and operations necessary to implementthe described implementations can also be stored on the machine-readablemedium. The instructions stored on the machine-readable medium can beexecuted by a processor or other suitable processing device, and caninterface with circuitry to perform the described tasks.

The above-described embodiments are intended to be examples only.Alterations, modifications and variations can be effected to theparticular embodiments by those of skill in the art without departingfrom the scope, which is defined solely by the claims appended hereto.

What is claimed is:
 1. A clock generator unit to generate a target clocksignal having a target frequency, the clock generator unit comprising: aFractional-N phase-locked loop (FNPLL) circuit having: first phasedetector circuitry having a FNPLL reference input terminal and a FNPLLfeedback input terminal, the FNPLL reference input terminal to receive afirst clock signal, the first clock signal having associated theretofirst clock signal Phase Noise; first voltage-controlled oscillator(VCO) circuitry operationally connected to the first phase detectorcircuitry, the first VCO circuitry to receive an output from the firstphase detector circuitry; mixer circuitry to receive a second clocksignal and an output signal from the first VCO circuitry, the secondclock signal being unrelated to the first clock signal, the mixercircuitry to generate a mixer output signal that comprises a sum signaland a difference signal, the sum signal being at a sum frequency that isa function of a sum of the frequency of the second clock signal and thefrequency of the output signal of the first VCO circuitry, thedifference signal being at a difference frequency that is a function adifference between the frequency of the second clock signal and thefrequency of the output signal of the first VCO circuitry; and filtercircuitry operationally connected to the mixer circuitry, the filtercircuitry to receive the mixer output signal and to generate anintermediate signal that is a function of the sum signal, the FNPLLcircuit configured to have a FNPLL bandwidth that filters out the firstclock signal Phase Noise and to generate the intermediate clock signalwith the intermediate clock signal being substantially free of the firstclock signal Phase Noise, and an Integer-N PLL (INPLL) circuit havingsecond phase detector circuitry and second VCO circuitry operationallyconnected to the second phase detector circuitry, the second phasedetector circuitry having an INPLL reference input terminal and an INPLLfeedback input terminal, the INPLL reference input terminal to receivethe intermediate clock signal, the second VCO circuitry to output thetarget clock signal, the INPLL circuit being configured to have an INPLLbandwidth that filters out Phase Noise generated by the second VCOcircuitry, the INPLL feedback input terminal configured to receive anINPLL feedback signal that is a function of the target clock signal, theFNPLL feedback input terminal configured to receive a FNPLL feedbacksignal that is also function of the target clock signal.
 2. The clockgenerator unit of claim 1 wherein the FNPLL circuit further hasFractional-N divider circuitry configured to receive, from the secondVCO circuitry, a signal that is a function of the target clock signal,the divider circuitry to generate the FNPLL feedback signal, a frequencyof the target clock signal being determined by at least settings of theFractional-N divider circuitry.
 3. The clock generator unit of claim 1wherein: the first VCO circuitry has a VCO and a VCO divider, the VCOdivider to receive a VCO signal from the VCO and to frequency-divide theVCO signal to obtain a frequency-divided VCO signal, the VCO signalhaving VCO Phase Noise, the frequency-divided VCO signal having lowerdBc/Hz Phase Noise than the first VCO signal.
 4. The clock generatorunit of claim 1 further comprising a clock source to generate the secondclock signal.
 5. The clock generator unit of claim 4 wherein the clocksource is one of a bulk-acoustic-wave resonator clock source, a microelectro-mechanical system resonator clock source, a surface-acousticwave resonator clock source and a quartz clock source.
 6. The clockgenerator unit of claim 5 wherein the Fractional-N PLL circuit, theInteger-N PLL circuit and the clock source are packaged as a unit. 7.The clock generator unit of claim 1 wherein: the first phase detectorcircuitry further has a prescaler divider and a phase detector, theprescaler divider having an input terminal to receive the first clocksignal, the input terminal of the prescaler divider being the FNPLLreference input terminal, the phase detector having a phase detectorreference input terminal, the prescaler divider to output a frequencydivided first clock signal to the phase detector reference inputterminal.
 8. A clock generator unit to generate a target clock signalhaving a target frequency, the target clock signal having a frequencyaccuracy that is a function of a frequency accuracy of a first clocksignal, the target clock signal having associated thereto target clocksignal phase noise that is a function of phase noise of a second clocksignal, the clock generator unit comprising: a Fractional-N phase-lockedloop (FNPLL) circuit having: first phase detector circuitry having aFNPLL reference input terminal and a FNPLL feedback input terminal, theFNPLL reference input terminal to receive a first clock signal, thefirst clock signal having associated thereto first clock signal PhaseNoise and a first clock signal frequency; first voltage-controlledoscillator (VCO) circuitry operationally connected to the first phasedetector circuitry, the first VCO circuitry to receive an output fromthe first phase detector circuitry; mixer circuitry to receive a secondclock signal and an output signal from the first VCO circuitry, thesecond clock signal being unrelated to the first clock signal, the mixercircuitry to generate a mixer output signal that comprises a sum signaland a difference signal, the sum signal being at a sum frequency that isa function of a sum of the frequency of the second clock signal and thefrequency of the output signal of the first VCO circuitry, thedifference signal being at a difference frequency that is a function adifference between the frequency of the second clock signal and thefrequency of the output signal of the first VCO circuitry; and filtercircuitry operationally connected to the mixer circuitry, the filtercircuitry to receive the mixer output signal and to generate anintermediate signal that is a function of the sum signal, the FNPLLcircuit configured to have a FNPLL bandwidth that filters out the firstclock signal Phase Noise and to generate the intermediate clock signalwith the intermediate clock signal being substantially free of the firstclock signal Phase Noise, and an Integer-N PLL (INPLL) circuit havingsecond phase detector circuitry and second VCO circuitry operationallyconnected to the second phase detector circuitry, the second phasedetector circuitry having an INPLL reference input terminal and an INPLLfeedback input terminal, the INPLL reference input terminal to receivethe intermediate clock signal, the second VCO circuitry to output thetarget clock signal, the INPLL circuit being configured to have an INPLLbandwidth that filters out Phase Noise generated by the second VCOcircuitry, the INPLL feedback input terminal configured to receive anINPLL feedback signal that is a function of the target clock signal, theFNPLL feedback input terminal configured to receive a FNPLL feedbacksignal that is also function of the target clock signal.
 9. The clockgenerator unit of claim 8 wherein the FNPLL circuit further hasFractional-N divider circuitry configured to receive, from the secondVCO circuitry, a signal that is a function of the target clock signal,the divider circuitry to generate the FNPLL feedback signal, a frequencyof the target clock signal being determined by at least settings of theFractional-N divider circuitry.
 10. The clock generator unit of claim 8wherein: the first VCO circuitry has a VCO and a VCO divider, the VCOdivider to receive a VCO signal from the VCO and to frequency-divide theVCO signal to obtain a frequency-divided VCO signal, the VCO signalhaving VCO Phase Noise, the frequency-divided VCO signal having lowerdBc/Hz Phase Noise than the first VCO signal.
 11. The clock generatorunit of claim 8 further comprising a clock source to generate the secondclock signal.
 12. The clock generator unit of claim 11 wherein the clocksource is one of a bulk-acoustic-wave resonator clock source, a microelectro-mechanical system resonator clock source, a surface-acousticwave resonator clock source and a quartz clock source.
 13. The clockgenerator unit of claim 11 wherein the Fractional-N PLL circuit, theInteger-N PLL circuit and the clock source are packaged as a unit. 14.The clock generator unit of claim 8 wherein: the first phase detectorcircuitry further has a prescaler divider and a phase detector, theprescaler divider having an input terminal to receive the first clocksignal, the input terminal of the prescaler divider being the FNPLLreference input terminal, the phase detector having a phase detectorreference input terminal, the prescaler divider to output a frequencydivided first clock signal to the phase detector reference inputterminal.